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For the JFET of Fig. 4-5; VDD 20 V; R1 1 M; R2 15:7 M; RD 3 k; RS 2 k; iG % 0; IDSSO 5 mA, and Vp0 5 V (and is temperature-independent). (a) Find the exact value of IDQ at 1008C. (b) Use sensitivity analysis to predict IDQ at 1008C. Ans: a IDQ 1:82 mA; (b) IDQ 1:84 mA 5.48 Solve parts a and b of Problem 5.25 if RS 2 k; VGG 1 V, and all else remains unchanged. Ans: a The transfer bias line is drawn on Fig. 5-14: IDQ max % 2 mA; IDQ min % 1:1 mA; (b) VDSQ max % 6 V; VDSQ min % 10:05 V
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Small-Signal Midfrequency BJT Ampli ers
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6.1. INTRODUCTION For su ciently small emitter-collector voltage and current excursions about the quiescent point (small signals), the BJT is considered linear; it may then be replaced with any of several two-port networks of impedances and controlled sources (called small-signal equivalent-circuit models), to which standard network analysis methods are applicable. Moreover, there is a range of signal frequencies which are large enough so that coupling or bypass capacitors (see Section 3.7) can be considered short circuits, yet low enough so that inherent capacitive reactances associated with BJTs can be considered open circuits. In this chapter, all BJT voltage and current signals are assumed to be in this midfrequency range. In practice, the design of small-signal ampli ers is divided into two parts: (1) setting the dc bias or Q point (s 3 and 5), and (2) determining voltage- or current-gain ratios and impedance values at signal frequencies.
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General hybrid-parameter analysis of two-port networks was introduced in Section 1.7. Actually, di erent sets of h parameters are de ned, depending on which element of the transistor (E, B, or C) shares a common point with the ampli er input and output terminals. Common-Emitter Transistor Connection From Fig. 3-3(b) and (c), we see that if iC and vBE are taken as dependent variables in the CE transistor con guration, then vBE f1 iB ; vCE iC f2 iB ; vCE 163
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Copyright 2002, 1988 by The McGraw-Hill Companies, Inc. Click Here for Terms of Use.
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If the total emitter-to-base voltage vBE goes through only small excursions (ac signals) about the Q point, then vBE vbe ; iC ic , and so on. Therefore, after applying the chain rule to (6.1) and (6.2), we have, respectively, @v @v vbe vBE % dvBE BE ib BE vce 6:3 @iB Q @vCE Q @i @i ic iC % diC C ib C vce 6:4 @i @v
B Q CE Q
The four partial derivatives, evaluated at the Q point, that occur in (6.3) and (6.4) are called CE hybrid parameters and are denoted as follows: @v vBE Input resistance hie  BE % 6:5 @iB Q iB Q @v vBE Reverse voltage ratio hre  BE % 6:6 @vCE Q vCE Q @i i 6:7 Forward current gain hfe  C % C @iB Q iB Q @i @ iC Output admittance hoe  C % 6:8 @v v
CE Q CE Q
The equivalent circuit for (6.3) and (6.4) is shown in Fig. 6-1(a). The circuit is valid for use with signals whose excursion about the Q point is su ciently small so that the h parameters may be treated as constants.
B + +
hre Lce _ E
hfe ib
hoe (S)
(a) CE small-signal equivalent circuit
ie + hrb Lcb _ B (b) CB small-signal equivalent circuit hfb ie hob (S)
Fig. 6-1
Common-Base Transistor Connection If vEB and iC are taken as the dependent variables for the CB transistor characteristics of Fig. 3-2(b) and (c), then, as in the CE case, equations can be found speci cally for small excursions about the Q point. The results are veb hib ie hrb vcb 6:9 ic hfb ie hob vcb 6:10
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